The present invention relates to secondary-side synchronous rectification in converters and more particularly to lessening conduction losses in rectifiers.
The incremental demand for size and weight reduction of portable and on board power modules has spurred significant development and research efforts in high-density, low profile power supplies. With ramping power density, thermal management becomes extremely critical and vital to the product quality and reliability. In the case of external adapters/chargers, designed for portable electronic equipment, that are housed in completely sealed enclosures for safety reasons, effective heat removal from the sealed box poses a major design challenge since the system cooling now primarily relies only on natural convection and radiation.
Resonant converters have become an attractive topology for front end DC bus conversion in medium and high power AC-DC power supplies. They offer significant advantages when compared to PWM converters, including soft-switching independent of load, reduced peak currents, higher power density, i.e., higher switching frequency making possible reduction in filter size, reduced electrical stress on power devices, and reduced EMI. Additionally, all these benefits are achieved without adding substantially to the overall system cost. The circuit parasitics can be used as energy storing elements of the resonant tank.
Among the different resonant topologies, LLC resonant converters have been the most attractive topology for front end DC bus conversion, as described in B. Yang, F. C. Lee, A. J. Zhang, G. Huang, “LLC resonant converter for front end DC/DC Conversion,” IEEE-APEC 2002, pp. 1108-1112. They utilize the magnetizing inductance of the transformer to construct a complex resonant tank with buck boost transfer characteristics in the soft-switching region, as described in Bo Yang, “Topology Investigation for Front End DC/DC Power Conversion for Distributed Power System” Dissertation, Virginia Tech, Blacksburg, Va., September 2003. (“Reference 1”).
Besides a few exceptions, described in Gerry Moschopoulos and Praveen Jain, “A Series Resonant DC/DC Converter with Asymmetrical PWM and Synchronous Rectification,” IEEE Transactions on Power Electronics, Vol. 3, pp 174-182, April 198; and J. A. Cobos, J. Sebastian, J. Uceda, E. de La Cruz, and J. M. Gras “Study of Applicability of Self-driven Synchronous Rectification to Resonant Topologies” Power Electronics Specialists Conference, 1992, pp 933-940, (“Reference 2”), there have been very few research efforts for implementing output synchronous rectification in the resonant converters. Due to the complexity, cost, and unsatisfactory performance of the existing discrete/integrated solutions, output rectification in these converters has been widely implemented using diodes.
Additionally, to meet the stringent standby power constraints from various Energy Compliance Agencies, e.g., Energy Star, CEC, etc., these converters are required to operate in burst mode during light load conditions. Existing controller integrated circuits (ICs) for synchronous rectification are based on Phase-Locked Loop (PLL) control and rely on synchronizing signals from the primary-side to anticipate the secondary device turn-off transitions. Apart from the fact that these control techniques are not suitable for synchronous rectification in resonant converters, these controller ICs cannot operate during burst mode conditions. Further, as described in Reference 2, the complex operating modes of resonant converters do not allow self-driven gate-drive techniques or the use of primary gate signals to drive the secondary-side MOSFETs.
High conversion efficiency, capacitive output filter, and reduced stresses across the output rectifiers make a half-bridge LLC series resonant converter an attractive topology for high voltage DC bus conversion. Additionally, the buck-boost transfer characteristics of such converter simplify design constraints for meeting system hold up time requirements and allows design of wide input range DC-DC converters without compromising on the converter performance. That is, because the system hold-up time specifications require the down converter to be designed for a wide input range operation. For conventional PWM converters, they can only be optimized at a lowest input voltage where the duty cycle reaches its maximum value and the efficiency drops at a high line where the duty cycle is much smaller. Therefore, the wide range operation dramatically reduces the converter efficiency at the normal operating conditions and increases the thermal management requirements for the system.
FIG. 10 shows a prior art dual output voltage resonant converter employing primary side switches M1 and M2, transformer T having two secondaries and standard Schottky diodes providing secondary side non-synchronous diode rectification. As is known to those of skill in the art, the design is simple and requires no secondary side controller, but the forward conduction losses of the diodes are substantial, particularly when the output voltages are low.
FIG. 1a shows a single output voltage series resonant converter 1, wherein the diodes have been replaced by controlled MOSFETs S1 and S2. The invention will be applied to control the secondary side switches S1 and S2 via SR (Synchronous Rectifier) controller 15. The converter includes a half bridge switching stage having high- and low-side switches M1 and M2 series connected at a switching node S. Gates of the high- and low-side switches M1 and M2 are connected to and driven by a controller IC 5 via resistors R1 and R2. The high-side switch M1 is connected to a voltage supply VIN and the low-side switch M2 is grounded.
The converter 1 further includes a transformer 10 having a primary coil LP series connected between an inductor Lr, which is connected to the switching node S, and a grounded capacitor Cr and parallel connected to an inductor Lm. Secondary coil portions LS1 and LS2 of the transformer 10 are series connected at a node P and to switches S1 and S2, respectively. Further, a parallel connected load RL and a capacitor Cout are coupled between the node P and the ground.
FIG. 1b illustrates typical waveforms made by the resonant converter 1. These waveforms include signals LO and HO produced by the controller IC 5 to drive the high- and low-side primary side switches M1 and M2; the voltage VP across the primary coil LP; the current IR through the inductor Lr, and the current ID from the secondary coil LS and through the switch S1.
By varying the switching frequency of the high- and low-side switches M1 and M2, which operate at 50% duty cycle, the controller IC 5 regulates the output voltage of the resonant converter. Tight output voltage regulation can be achieved by varying the switching frequency of the converter. The output voltage is regulated with respect to the frequency of the resonant tank formed by the inductor Lr, capacitor Cr, and the magnetizing inductance of the primary coil Lp of the transformer 10. The voltage at the primary coil Lp of the transformer 10 has a 50% duty cycle under all operating conditions and is phase shifted with respect to the voltage at the input of the resonant tank. Hence, control driven rectification cannot be implemented to drive the secondary devices.
FIG. 1c illustrates AC transfer characteristics for the converter 1. The converter has two resonant frequencies, a lower resonant frequency, provided by the inductors Lm, and Lr, the capacitor Cr and the load RL, and a higher fixed series resonant frequency fR1 provided by the inductor Lr and the capacitor Cr only. Hence, the two switch devices M1 and M2 can be soft-switched by operating the converter either above or below the series resonant frequency fR1.
Secondary-side synchronous rectification has widely replaced traditional diode based rectification implementations in a number of low voltage DC-DC converter applications. Due to design simplicity, two control schemes have been widely adopted for implementing output synchronous rectification. The first control scheme to derive the gate drive signals for the secondary side MOSFETs is control driven rectification using a secondary controller/driver IC with synchronizing signals from the primary gate-drive. However, deriving the optimum secondary gate-drive signals still presents a significant challenge in a number of Switched-Mode Power Supply (SMPS) topologies. The other control scheme is self-driven rectification using signals directly from the output of the power transformer. Here, an output of the power transformer is used to drive the secondary devices.
FIG. 2a illustrates different operating modes of the series resonant converter 1 of FIG. 1a. The different operating modes or load conditions depend on the level of the load current IOUT. The top graph in each mode illustrates the midpoint of the half bridge and voltage across the resonant capacitors. The middle graph illustrates current through the resonant inductor and magnetizing inductance. The bottom graph illustrates the output current IO. Where mode A IOUT=I1 is a nominal operation condition known as the Continuous Conduction Mode (CCM) because the output current is always continuous, where the primary coil Lp is clamped by the output voltage and never operates in the resonant process. As the load current is decreased, the converter changes to operation in Mode B, IOUT=I2 and the resonant tank consists of the capacitor Cr and primary coil Lp+the inductor Lr. There is a small time period when the secondary current IO is zero (the primary current is clamped to the magnetizing current) after the primary devices switch. As the load is decreased further, the converter changes to operation in Mode C, IOUT=I3, where I3<I2<I1, where the tank current is now clamped to the magnetizing current even before the primary switching devices M1 and M2 switch.
In the case of resonant converters, the output synchronous rectifiers S1 and S2 cannot be driven by using synchronizing signals from the primary-side due to the phase shift between the resonant tank voltage and current as observed in FIG. 2b. The top graph of FIG. 2b shows the phase of the primary-side gate signal waveforms and the bottom graph shows the phase of the secondary rectifier voltage and current waveforms lagging behind. The relative phase shift varies depending on the operating point of the converter with respect to the resonant frequency of the LLC tank.
Self-driven synchronous rectification for various resonant converter topologies has been investigated, as discussed in Reference 2. For the LLC resonant converter, the current through the output rectifiers behaves differently depending on the operating point of the converter with respect to fR1, as described in Reference 1. When the converter operates at light load conditions in the region above fR1, the output current IO is discontinuous i.e. the secondary winding voltage and currents are not in phase as shown in FIG. 2c. Waveforms during operation at light load (f>fR1)—voltage across the switch S2 and current through the switch S1 illustrated at the top of the drawing and discontinuous output current IO illustrated at the bottom.
Hence, self driven rectification cannot be implemented for converters operating in this region above fR1 as the timing mismatch would cause Cout to discharge across the ON rectifier. This results in a reactive power flow between the output and the power transformer is shown in FIG. 3. Reactive power flows between the output and power transformer. The graph (a) shows the resonant tank current; the graph (b) shows the secondary rectifier current through the secondary switch; and the graphs (c) and (d) gate-source and drain-source switch voltages VGS and VDS. Hence, as illustrated, self driven rectification cannot be implemented here due to the discontinuity in the secondary current IO, i.e., the output capacitor Cout will discharge across the ON secondary switch when that device's current is zero.
Similar load dependent properties of other resonant converter topologies add complexity in implementing synchronous rectification at the output. When the converter operates in the soft switching range below fR1, the magnetizing inductance participates in the resonant cycle and results in discontinuities in the output current. This is shown in FIG. 4, where waveforms for operation at f<fR1, voltage and current across the primary of the power transformer and output current IO. Hence, self driven rectification cannot be implemented when operating in this region as well.
As discussed above, no dedicated IC is provided that is targeted towards synchronous rectification in resonant converter applications. On the other hand, implementing synchronous rectification in these topologies with discrete control is cumbersome and requires at least two current sense transformers, two high speed, high voltage comparators, and a high current, low propagation delay gate-driver to drive the two rectifiers. Additionally, large printed circuit board (PCB) area will be required for this approach, and the discrete component tolerances and variations across temperature range strongly affect timing performance, which is extremely critical here.
What is needed is a control that operates completely independent of the primary side control. It should be based on differentially sensing the drain-source voltage of the MOSFET device to determine the level of current through the MOSFET device and turn the MOSFET device ON and OFF in close proximity of the zero current transition; hence, the control should always ensures zero-voltage turn on transitions. The sensed voltage should be compared with two negative thresholds to decide the turn-ON and turn-OFF transitions for the synchronous rectifiers.